Waveguide filter

ABSTRACT

This invention relates to waveguide filters wherein waveguide sections functioning in the evanescent mode for frequencies in the desired passband are utilized to couple conventional cavity filter sections together. The evanescent sections operate in their normal modes at frequencies higher than the passband frequencies. Suppression devices are then coupled to one or more of the evanescent sections (instead of to the cavity filter sections) to suppress the parasitic harmonic waves while having negligible effect on the evanescent mode passband frequencies.

United States Patent [72] Inventor George Frederick CravenSawbridgeworth, England [21] Appl. No. 671,046

[22] Filed [45] Patented [73] Assignee Sept. 27, 1967 Jan. 11, 1972International Standard Electric Corporation New York, N.Y.

[54] WAVEGUIDE FILTER 4 Claims, 44 Drawing Figs.

[52] U.S. Cl 333/73, 333/83, 333/81 [51] Int. Cl ..I-l03h 13/00, H03h7/10 [50] Field of Search 333/73, 73 C, 73 W, 10, 7, 9

[56] References Cited UNITED STATES PATENTS 3,215,955 11/1965 Thomas333/7 2,866,595 12/1958 Marie... 333/10 2,849,683 8/1958 Miller 333/103,058,072 10/1962 Rizzi 333/73 W 2,626,990 l/1953 Pierce 333/9 2,623,12012/1952 Zobel .1 3113/73 W 2,816,270 12/1957 Lewis 1133/) 2,106,7682/1938 Southworth. 333/73 W 2,866,949 12/1958 Tillotson 333/11 3,451,0146/1969 Brosnahan... 333/73 3,237,134 2/1966 Price 333/73 W PrimaryExaminer-Herman Karl Saalbach Assistant Examiner-C. Baraff Attorneys-C.Cornell Remsen, .lr., Rayson P. Morris, Percy P. Lantzy, Philip M.Bolton and Isidore Togut ABSTRACT: This invention relates to waveguidefilters wherein waveguide sections functioning in the evanescent modefor frequencies in the desired passband are utilized to coupleconventional cavity filter sections together. The evanescent sectionsoperate in their normal modes at frequencies higher than the passbandfrequencies. Suppression devices are then coupled to one or more of theevanescent sections (instead of to the cavity filter sections) tosuppress the parasitic harmonic waves while having negligible effect onthe evanescent mode passband frequencies.

PATENTEU JAN] 1 I972 SHEET OlUF 15 1- JZO/ 1 591M Inventor GEORGE F.CRAVEN Agent PATENTED JAN! 1 1972 3534788 SHEET OBUF 15 iii F //MInventor GEORGE E CRAVEN Byww Agent PATENTEI] JMI I I H12 SHEET USUF 15Inventor GORG F. CRAVEN vi 74? Agent PATENTEUJmmsm 3634-788 SHEET 080F15Fmguencyfl/cp In ventor GEORGE E CHM/51v Agent PATENTEI] mu 1 i972 SHEETOSUF 15 I nvenlor Geeks: F. CRAVEN By Agent PATENTED JAN] 1 i97231634788 SHEET IUUF 15 v L l I l lnvenlor GEORGE F. CRAVEN y Agent ammaPATENTEI] JAN! 1 i972 sum 11 0F 15 gam 0000 Fnequenqy Mc/S InventorGEORGE F. CRAVEN Agent PATENTED mu 1 1912 315347 saw 12 0F 15 FrequenzyFrequency QM- lnvcnlor GEORGE F. CRAVN Agent PATENTED mu 1 m2 3.634.788

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Inventor GEORGE F. (RAVEN By Agent WAVEGUIDE FILTER BACKGROUND OF THEINVENTION This invention relates to waveguide band-pass filters.

Conventional direct-coupled and quarter-wave-coupled filters each haveadvantages and disadvantages. The advantages of the quarter-wave-coupledfilter lies in its small coupling susceptances and consequently, theease with which a paper design can be realized with practical mechanicaltolerances. Initially, this type of filter was preferred because thecavities could be made separately and tuned before assembly. However,now that filters are always made and tuned as a complete unit thisadvantage is of little value and general, if not complete, preferencehas passed tothe direct-coupled filter. The direct-coupled filtereliminates the frequency-sensitive quarter-wave couplings and,therefore, is applicable to wider bandwidth designs; it is also shorter(about 75 percent of the length of the quarter-wave filter). This isachieved by substituting one large susceptance for the two smallersusceptances and the quarter-wave coupling between cavities. This makesthe direct-coupled filter potentially cheaper although this is offset bythe much stricter tolerances, or additional adjustments, that arenecessary. The difficulties arise from the highly critical nature oflarge susceptances which, in the example of a symmetrical diaphragm, forinstance, varies as cot (1rd,/2a)(d,1 for large susceptances). Use hasalso been made of multipost arrangements but this alternative leads tomanufacturing difficulties and the thin posts also increase loss.

SUMMARY OF THE INVENTION Therefore, the main object of this invention isto provide an improved waveguide filter wherein the higher orderparasitic frequencies are suppressed while having negligible effects onthe desired passband signals.

According to the invention there is provided a waveguide band-passfilter having main resonator cavities coupled together by waveguidesections which are evanescent at the passband frequency of the filter.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1(a) is the equivalent circuit oftwo cavities with a conventional quarter-wave coupling,

FIG. 1(b) is the equivalent circuit of two cavities with a conventionaldirect coupling,

FIG. 1(c) is the equivalent circuit of two cavities with an evanescentmode coupling,

FIG. 1(d) is the voltage transfer equivalent of an evanescent section,

FIG. 2(a) is the bisected circuit of FIG. 1(a).

FIG. 2(b) is the bisected circuit of FIG. 1(c).

FIG. 3 shows a typical attenuation/length characteristic for anevanescent mode coupling measured at 4,000 mc./sec.,

FIG. 4 shows the evanescent mode coupling giving the characteristic ofFIG. 3,

FIGS. 5(a) and 5(b) show a symmetrical evanescent mode coupling itsequivalent circuit, respectively,

FIG. 6 shows the reactance at a junction of a waveguide terminated inthe symmetrical evanescent section of FIG. 5(a).

FIG. 7(a) and 7(b) show an asymmetrical evanescent mode coupling and itsequivalent circuit, respectively,

FIG. 8 shows the reactance at a junction of a waveguide terminated inthe symmetrical evanescent section of FIG. 7(a),

FIGS. 9(a), 9(b) and 9(0) are plan, side and end views, respectively, ofa six-section resonant cavity waveguide bandpass filter with evanescentmode coupling sections,

FIG. 10 is a perspective view, partially cut away, of a modified form ofan evanescent mode coupling section,

FIG. 11 shows the curves insertion loss and V.S.W.R. vs. frequency forthe filter of FIG. 9,

FIG. 12 shows the measured standing wave pattern in a three-sectionfilter with evanescent mode coupling sections throughout,

FIG. 13 shows the bisected equivalent circuit of a symmetrical networkof imaginary characteristic impedance terminated for full energytransfer,

FIG. 14 shows the calculated standing wave pattern for the symmetricalhalf-section evanescent coupling between cavities of FIG. 13,

FIGS. 15(a) and 15(b) are plan and sectioned side views, respectively,of an evanescent coupling section with a harmonic rejection filter,

FIGS. 16(a) and 16(b) are plan and sectioned side views respectively ofan evanescent coupling section with a constant resistance rejectionfilter,

FIG. 17 shows the transmission response of the filter of FIG. 16,

FIG. 18 is a perspective partially cutaway view of an evanescentcoupling section with a resonant slot hybrid junction rejection filter,

FIG. 19 shows the attenuation characteristic of the filter of FIG. 18,

FIG. 20 is a perspective partially cutaway view of an evanescentcoupling section with a form of constant resistance rejection filter,

FIG. 21 shows the attenuation characteristic of the filter of FIG. 20,

FIG. 22 shows the equivalent circuit of the filter of FIG. 20,

FIGS. 23(a) and 23(1)) are side and end views, respectively, of anevanescent coupling section with a series resonant rejection filter,

FIG. 24 shows the attenuation characteristic of the filter of FIG. 23,

FIG. 25 is a perspective partially cutaway view of an evanescentcoupling section with a form absorption filter,

FIG. 26 shows the standing wave pattern in the coupling section of FIG.25,

FIG. 27(a) is an end view, and FIG. 27(b) is a half-sectioned side viewalong the line A-A of FIG. 27(a), of an evanescent coupling section witha low-pass rejection filter,

FIG. 28 shows the attenuation characteristic of the filter of FIG. 27,

FIG. 29 shows a typical frequency characteristic of a waveguide low-passfilter,

FIG. 30 shows the wide-band rejection characteristics of a waveguidelow-pass filter, and

FIG. 31(a) is a plan view, and FIG. 31(b) a section on the line 8-8 ofFIG. 31(b), of cascaded low-pass filter evanescent coupling sectionswith a dissipative section in between.

DESCRIPTION OF THE PREFERRED EMBODIMENTS Waveguide at frequencies belowcutoff exhibits characteristics common to all nondissipative filternetworks in their stop-band region. The characteristic impedance, whichis real in the passband becomes imaginary in the stop-band. Thepropagation constant, which is imaginary in the passband becomes real inthe stop-band. The transmission line analogue then has the propertiesshown in FIG. I(c), FIGS. 1(a) and I(b) represent the correspondingcircuits for quarter-wave and direct-coupling, respectively. Theconditions for virtual identity between the two circuits of FIGS. 1(a)and 1(b) have been established for example, in An Improved DesignProcedure of the Multisection Generalized Microwave Filter," Levy R.,I.E.E. Monograph Vol. 232R, Apr. 1957, in which it is shown that if theinsertion loss and phase shift of the two networks are the same, thecircuit of FIG. 1(a) may be replaced by that in FIG. 1(b). In a similarway the network of FIG. 1(c) may be replaced by that in FIG. 1(b). Theinput reactance of the network of FIG. 1(b) is equivalent to a certainline length in the network of FIG. 1(c) and thus, the cavity length willbe generally less when evanescent couplings are used. The insertion lossof the two networks may then be made identical by appropriate choice ofthe parameters in the evanescent section.

The conditions for equivalence between the two networks of FIGS. l(b)and 1(c) can be readily derived from the standard transmission lineequations. As the networks are symmetrical they may be bisected (FIGS.2(a) and 2(b) nd the openand short-circuit admittances derived. If therespective open circuit and short-circuit admittances can be equated andsolved for physically realizable values then the two networks may bemade identical at a given frequency. The short-circuit admittances forthe networks of FIGS. 2(a) and 2(b) are, respectively.

Y,,,.,=j Yo cot I) Y,,,. =jB j m" coth (pl/2 The open-circuitadmittances for the same networks are:

'M/=1 (v+ (3) where d) cot (B ,2Yv)

Y =jB,jY0i tanh (Ill/2) Equating the short-circuit admittances.

Y0 cot 0=B,+Y0i coth (111/2) (5) Equating the open-circuit admittances.

Y0 cot (0+)=B,+Yoi tanh (Ill/2) Expanding the left hand side (6) andsubstituting for cot 6 (from (5)) we have, solving for cot 0,

B Yoi B Yet 71 B Yoi 71 Got Yet h 1 h w To (cot tan The extreme valuesthat cot can have are when 71- wand l 0. In the first case both coth andtanh g tend to unity. Thus, cot w as the line length tends to infinitywhich is the expected result. In the second case as 'yl 0 the expressionreduces to:

which is obviously true. Equation (5) states simply the secondrequirement for equivalence i.e., the relationship between the linelength in FIG. 2(a) and the combined lumped susceptance including thejunction susceptance and the (inductive) characteristic admittance oftheline.

The notion of energy propagating freely through a nondissipative line(or guide) of imaginary characteristic impedance, with the propagationconstant real and assuming large values, may conflict with the intuitivenotions held by many. This con cept amounts to the belief that the ratioof input voltage, E to output voltage, E is given to good accuracy bythe equation:

(E,/E )=ev where y propagation constant I =length However, this is onlytrue if the network is terminated in its characteristic impedance(imaginary). With other terminations the results can be very different.For instance, if the termination is capacitive (FIG. 1d) the ratio E /Eis given by For large values of y], cosh 71 and sinh y! are nearlyequal. Thus, for example, if Z0 -x E /E is quite small and may be zero.The circuit then behaves as a series resonant circuit with a largeresonant rise in voltage across the inductance and capacitance. Clearly,a wide range of values including E,/E =l is obtainable. A generalengineering conception of the behavior is simply to say that thereactive insertion loss of evanescent guide may be tuned out" by asuitable termination. Such evanescent waveguide sections are generallydescribed on my copending application, Ser. No. 643,279. It roughlycorresponds to neutralizing the effect of a shunt susceptance by addingone of equal value (but opposite sign) in shunt with it.

The design of direct-coupled filters is described in Direct CoupledFilters, Cohn S. B., Proc. I.R.E., Feb. 1957. The

which for large values of b is, to good accuracy,

10 Iog The insertion loss of a section of evanescent guide interposedbetween matched guides will be a function of the guide length andjunction susceptance. The guide length will normally be the principalfactor, the insertion loss being:

and M cutotf wavelength A0 free space wavelength.

The junction susceptance is generally quite large but the insertion lossfrom this effect is less than might be expected. The reason for this isthat the field from the junction spreads into the evanescent sectioncausing the initial rate of decay to be less than e1 FIG. 3 shows theinsertion loss, as a function of length, of a typical section shown inFIG. 4 of l.00OXO.667- inch waveguide 41 coupling 2.000XO.667-inchwaveguide 42. Except when the evanescent section is very short (andultimately degenerates into a thin iris) the curve is a straight line.Thus, only two properly chosen measurements are essential in order toestablish a design curve. Comparing the calculated (eq. (10)) andmeasured curves it will be seen that they are approximately 2 db. apartand parallel within about I db. over the range. Therefore, eq. (10) willgive a very accurate guide to the effect that minor variations in A,will have on the measured attenuation curve. It is of interest that thedifference between the two curves is roughly one-half the insertion lossof an iris of the same dimensions as the cutoff guide. This is a usefulrule of thumb correction, if it is desired to use the theoretical curvefor ratios a/a (a'la is defined in FIGS. 5 to 8).

The practical design of a filter is carried out in the following way.The values of the prototype sections are calculated using the equationsin FIG. 2 of Cohn's paper. The susceptance values are then determinedfrom FIG. 5 of the same paper. The insertion loss is then calculatedfrom eq. (9) of this specification. A width dimension, which is wellbeyond cutoff at the passband frequency is chosen, and the inputreactance of the section is then determined (FIG. 6 or 8). Thehalflength of the cavity is then found from:

Where tan (l,/Z0) (see FIG. 6 or8 for value of(x/Zo). If the cavity isphysically symmetrical about the electrical center i.e., the inputreactance at each end is identical) the total length will be twice thevalue found in eq. (ll); otherwise the second electrical length iscalculated. If the total cavity length is unreasonably short as aconsequence of the end loading, a second trial using a smaller guidewidth for the evanescent section will be necessary. The length of thesection which will yield the desired insertion loss is then found from acurve such as that shown in FIG. 3.

Where the precise value of the filter bandwidth is not very important anapproximate design technique using eq. 10) with an approximatecorrection for the discontinuity, can be employed. The desiredcharacteristic maximally flat or Tchebycheffwill be obtained even if thetheoretical attenuawhere tion curve is in error. This follows becausethe correct ratio between the insertion lossvalues will be realized. Theoverall I error in bandwidth can be gauged roughly by noting that anerror of 0.5 db. in all of the intercavity coupling sections willproduce an error of about 5 percent in the bandwidth of a typicalfilter.

So far only the intercavity couplings have been discussed as it is inthese sections that evanescent couplings have their chief application.Susceptances at the end of the filter are usually of a comparatively lowvalue and may, conveniently, be of the conventional type. However, whereit is desirable to preserve the same type of construction throughout thefilter it is possible to use evanescent sections at the end.

FIGS. 9(0), 9(1)) and 9(0) show a six-section maximally flatdirect-coupled filter with a 3 db. bandwidth of 37 mc./sec. The 3 db.limit frequencies are 3,983 mc./sec. and 4,020 mc./sec. the constructionbeing in 2=2/3 in guide. There are evanescent mode coupling sections 1between the resonant cavities 2. Each cavity 2 has a 2-BA coarse-tuningscrew 3 and an 8-BA fine-tuning screw 4, and each evanescent section 1has A O-BA coupling adjustment screws 5. Inductive arises 6 at the endof the filter are conventional.

The physical dimensions of the filter are given below:

The first step in designing the filter was to calculate the values ofthe prototype sections using Cohns paper. For a maximally flat filter,using the same notation as Cohn, the values are:

The guide wavelength tenns A, and A are the values occurring at the 3db. band limits.

Taking x,,,=11.207 cm. and A =l0.981 cm. F0.032l. Substituting for L andg in 12) the values of B,, H, are:

The value of the outer susceptances B and 13,, are obtained bysubstituting g,,=L in 12) which gives The insertion loss of 8, B and Bare from (9):

L, =22.49 db=L L =28.23 db=L L ,=29.57 db.

The dimensions of the evanescent guide section may now be calculated.Choosing a symmetrical junction with guide widths of l in. (evanescentsection) and 2 in. (propagating section) respectively, the valueXZ,,/2a)t may be obtained from FIG. 6. For a'/a=0.5;)t,,=l1.l cm. X(Z,,normalized) is given by X=0.3l. From (II) the electrical half-length ofthe cavity or a total length, for identical junctions at each end of thecavity, of 145.6". This gives a value of l 145.6Xl I.l/360)=4.49 cm. (or1.77 in.)

Reducing this value by percent to allow for tuning screws we have:

The details of the inductive irises at the end of the filter remain tobe settled. These are' conventional symmetrical irises the dimensions ofwhich may be obtained from any standard text.

The performance of the filter is shown in FIG. 11.

The field distribution in evanescent modes conveying full power throughthe section is of interest. FIG. 12 shows the measured fielddistribution in a typical three-section filter using evanescentcouplings throughout. The filter was constructed with a slotted linesection and measurements made with a probe (with very small insertion)in the usual way. When measuring the field in the various sections itwas necessary to close up the slot in order to prevent radiation fromthe asymmetrical end sections. This was achieved by fitting a number oftongued sections (similar to the probe section) into the slot. Thesewere linked to each other and attached to the probe so that theytravelled along the guide with it.

FIG. 14 gives the calculated field along a transmission line ofimaginary characteristic impedance which is terminated for full energytransfer (HG. 13). The network is symmetrical and it is, therefore,sufficient to bisect the network and calculate E lE as a function of 1(equation (8)). The values of y and x, (x, is the equivalent reactancepresented by the cavity at the line terminals) are chosen for a typicalexample. It will be seen that a standing wave exists on the linereaching zero in the dissipationless case at the center andcomparatively small values if the loaded Q is high. The generalcharacter of results is the same at that shown in the experimentalresults of 12.

It is of interest that the magnitude of the electric field behaves inmuch the same way as if an iris of very high susceptance were located atthe center of the intercavity coupling section. The end sections behavedifferently owing to their asymmetrical nature. The semiresonant rise involtage predicted by eq. (8) can be clearly seen in FIG. 12.

The essential difference between this type of band-pass filter andconventional types lies in two features: the physical length of thecoupling sections and the nature of the field existing within thesections. The insertion loss of an evanescent section depends both onits width and length and, therefore, the length may vary over a widerange of values. This means that the filter will be somewhat longer thanthe conventional direct-coupled type although, in general, it need notbe longer than the quarter-wave coupled filter. This is a disadvantagewhere the shortest possible filter is desired. However, where a modestincrease in length can be tolerated the filter has a number ofadvantages.

a. The freedom of choice in the length of the coupling sections meansthat the overall length of a given type of filter can be maintainedirrespective of its center frequency. This follows, because of thelonger wavelengths the greater cavity length is obtained by reducing thelength (and if necessary, the width) of the evanescent section. Thus,only one fixed body size, with appropriate drillings, needs to beretained for a complete waveguide band. This simplified storage problemsin production. The inserts, which form the evanescent sections, and areconsiderably more robust than conventional irises, are then the onlyvariables.

b. The mechanical tolerances for large values of B, by whatever methodthat are obtained, are very severe. For example,

if the coupling secfions in the center of the filter of FIG. 9 are Ireplaced by symmetrical inductive irises of the appropriate value, thetolerance on the width dimension (i.e., the gap between the irises) isonly one-third of that which may be permitted with the correspondingevanescent guide dimension. Other types of obstacles, such as holes andmultipost arrangements require even stricter tolerances. Therefore,filters in which the bandwidth is significantly narrower than about Ipercent (the above example) will require considerably less severetolerances if evanescent coupling sections are employed.

c. The style of construction considered so far can be produced byinserting milled blocks in waveguide in order to produce the evanescentsections. Another method is to spray metal on a suitable mandrel. Thefilter can be produced in two halves by milling from solid material.This method of construction can be expected to yield very high precisionand has the advantage of eliminating soldered joints.

d. Variable bandwidth filters can be produced by using variable-cutoffevanescent coupling sections such as that shown in FIG. 10, whichincorporates a movable block 7 locked in any desired position by a screw8 and additional to fixed blocks 9 and 10.

e. The most important feature of this filter is its potential foreliminating the parasitic passbands that are characteristic of allband-pass filters at microwave frequencies. Normally, suppression ofthese passbands requires an additional filter which then affects themidband insertion loss and reflection. However, with evanescent couplingsections this disadvantage is avoidable. One or more of the couplingsections can be so designed that the frequency it is desired to suppresspropagates in the coupling section instead of being an evanescent wave,like the desired passband. It is then possible to insert parasiticsuppressors in the section concerned which will have a very large effecton a progressive wave, but none on an evanescent wave.

Conventional resonant-cavity waveguide band-pass filters suffer fromunwanted transmission bands above the desired passband. These passbandsoccur for two reasons. Firstly, in direct-coupled resonant-cavityfilters resonance occurs at frequencies for which the electrical lengthof the cavity is n. Ag/Z (where n is an integer and Ag is guidewavelength). In addition to these multiple resonances, which areapproximately in harmonic relationship, further resonances can beexpected as a result of higher order modes propagating. In general,individual cavities tend to resonate at slightly different frequenciesand interaction between these resonances occurs. As a result, multiplenarrow passbands occur at unpredictable frequencies and, therefore,there is no way of knowing whether a filter will provide satisfactoryharmonic rejection, for example, before it is built. If, in addition, aspecific filter design is to be tuned over a wide range of frequenciesthe danger of harmonics penetrating the filter becomes great. In thesecircumstances, where suppression of high-power harmonics is essential,it is necessary to cascade a second filter giving broad bandsuppression. This may take the form of a corrugated (or waffle-iron)low-pass filter or a leaky wall filter. The former is based on theclassical lumped circuit analogue and is a reflection filter; the secondpermits the harmonics to leak through the wall of the filter toauxiliary guides where they are dissipated. The difficulty with both ofthese filters is that they affect the passband performance adversely.Both filters add unavoidable dissipation and reflection that cannot bematched out (because the phase of the reflection varies rapidly withfrequency).

The behavior of filters using evanescent mode couplings is completelydifferent from conventional iris-type filters at frequencies will abovetype filters at frequencies well above the main transmission band. Abovea critical frequency the coupling section propagates and the resonantcavities formed by the evanescent coupling sections virtually disappear.Instead, the filter behaves, practically, as a straight through deviceof nearly zero attenuation and therefore, only has minor internalreflections. In this form it is worse than the conventional filter, butthe elimination of large internal reflections in the filter is aprerequisite to providing predictable suppression of unwantedfrequencies. The second necessary condition is that the couplingwaveguide should behave differently for the passband and unwantedfrequencies, respectively. In this way devices which affect the unwantedfrequency will have little or no effect on the desired frequency. Thebehavior of evanescent couplings, as described above, suits them to thisapplication.

In general, the suppression of unwanted frequencies presents itself intwo possible forms. One occurs when the filter functions in a closedcircuit e.g., it provides filtering in the output of an oscillator. Theunwanted frequencies, virtually entirely harmonics, are known. If thedevice is a highpower oscillator a very large amount of suppression maybe necessary, but it will be required at specific frequencies i.e.,narrow-band rejection circuits of very high rejection will be required.The second example occurs when the filter is located in an open-circuitsystem i.e., a system connected, for example, to a wide-band aerial.Signals over a wide band of frequencies will be likely to be receivedand, thus, wide-band suppression is necessary. In general, the degree ofsuppression required will be required to be less than the first example.

FIGS. 15(a) and 15(b) shows an evanescent coupling section 11 betweenresonant cavities 12 incorporating a narrowband harmonic rejectionfilter. At the desired transmission band the coupling section isdesigned to be evanescent but the dimensions are chosen so thatpropagation occurs at the frequency to be suppressed. A three-sectionrejection filter is shown, the stubs 13 containing semiconductor tuningpistons 14 being conveniently connected to the guide by the broadbandresonant slots 15. Approximately 35 db. rejection per section can beobtained in a typical example over a relatively narrow band. Thus, athree-section filter built in the coupling sections as shown givesrejection, at a given frequency, in excess of I00 db. If it is desirableto absorb the undesired harmonic, rather than reflect it back to thesource, the filter can be converted into a constant resistance filter(British Pat. No. 1,018,923 G. F. Craven 7) and the energy absorbed intothe matched load 16, FIGS. 16(a) and 16(b) in which like references havebeen used as for FIG. 15. The rejection of one section is largely lostbut the advantage of being able to absorb the energy is oftensignificant. The performance of a filter using one of the sections ofFIG. 16 is shown in FIG. 17. The midband insertion loss of this section(0.4 db.) compared with a section in which no rejection sections wereincluded was identical within the limits of measurement. The reason forthe absence of additional loss is that the guide section incorporatingthe load is beyond cutoff at the main transmission frequency and caneasily be made sufficiently long to prevent measurable loss. In thisrespect, this form of the rejection filter has something in common withleaky wall filters which dissipate the energy by coupling to a largenumber of guides (which are beyond cutoff at the main transmissionfrequency). However, this filter avoids the main disadvantage of theleaky wall filter: the main transmission frequency propagates in thedominant mode and therefore, harmonics propagate in several modes.Multiple coupling to all possible modes is then necessary and thisproduces measurable insertion loss at the main frequency. In manyapplications the present filter is likely to be superior to the leakywall filter, because the guide size of the coupling section may bechosen so that the harmonic to be suppressed propagates in the dominantmode only (the main frequency propagating in an evanescent mode). Thus,the present filter will be much simpler, have less insertion loss andcombine the functions of band-pass filter and harmonics suppressionfilter.

Since the filter of FIG. 15 is essentially a reflection filter, only onesuppression filter of this type is pennissable per complete band-passfilter. This is because, if additional evanescent coupling sections areconverted into suppression filters, mutual cancellation can be expectedwith further parasitic passbands resulting. Thus, the amount ofsuppression that can be obtained is limited to the number of suppressionsections that can be contained in one coupling section.

However, this limitation does not apply if the suppression filters aredissipative. Ordinarily, dissipation in the main coupling arm is notacceptable, but if the suppression filter is designed as a constantresistance type a considerable resistive attenuation is obtainable atthe parasitic frequency. At the desired passband frequency the couplingguide is evanescent and therefore no dissipation can occur.

- Another form of narrow-band suppression filter is shown in FIG. 18.Two resonant cavities 17 are coupled by an evanescent coupling sectionI8. Resonant coupling slots 19 .couple the evanescent section 18 to alength of circular

1. A waveguide band-pass filter comprising: at least one waveguidesection which is evanescent for the passband frequency of the filter,said waveguide section being dimensioned to propagate parasitic passbandfrequencies; at least a first and second resonator cavity filter sectioncoupled together through said one evanescent waveguide section; anarrow-band absorption filter coupled to said evanescent waveguidesection for suppressing parasitic frequencies, said absorption filter isof the constant resistance type and includes: at least two stubs; meansincluding at least two broadband resonant slots for coupling said atleast two stubs to said evanescent waveguide section; means coupled toone of said stubs for absorbing reflected energy from other of saidstubs coupled to said evanescent waveguide section; at least one tunablescrew coupled to the other of said at least two stubs.
 2. A waveguideband-pass filter comprising: at least one waveguide section which isevanescent for the passband frequency of the filter, said waveguidesection being dimensioned to propagate parasitic passband frequencies;at least a first and second resonator cavity filter section coupledtogether through sAid one evanescent waveguide section; a narrow-bandabsorption filter coupled to said evanescent waveguide section forsuppressing parasitic frequencies, said absorption filter includes: athird cavity resonant at the parasitic frequency; a fourth cavity; meansincluding a coupling hole in each of said third and fourth cavities totransfer energy at the parasitic frequency from said third cavity tosaid fourth cavity; a matched load coupled to said fourth cavity forabsorbing said transfer energy; and a series rejection filter sectioncoupling said third and fourth cavity to said evanescent waveguidesection.
 3. A waveguide band-pass filter comprising: at least onewaveguide section which is evanescent for the passband frequency of thefilter, said waveguide section being dimensioned to propagate parasiticpassband frequencies; at least a first and second resonator cavityfilter section coupled together through said one evanescent waveguidesection; a narrow-band absorption filter coupled to said evanescentwaveguide section for suppressing parasitic frequencies, said absorptionfilter includes: a third cavity resonant at the parasitic frequency; anobstruction mounted within said third cavity for absorbing the parasiticfrequency energy without having any significant effect on the passbandenergy; and a series rejection filter coupling said parasitic frequencycavity to said evanescent waveguide section.
 4. A filter according toclaim 3 wherein said obstruction includes an absorption screw.